Circuits and methods for controlling a charge pump system

ABSTRACT

A circuit includes a charge pump and a feedback circuit. The charge pump coupled to a switch provides a control signal to the switch. The feedback circuit coupled to the charge pump receives the control signal and adjusts an operating frequency of the charge pump based upon the control voltage. The control voltage is adjusted to a predetermined target voltage by adjusting the operating frequency through the feedback circuit.

RELATED APPLICATION

This application is a continuation-in-part of the co-pending U.S. patent application Ser. No. 12/313,114, titled “Charge Pump Systems with Adjustable Frequency Control,” filed on Nov. 17, 2008, which claims priority to U.S. Provisional Patent Application Ser. No. 61/003,998, filed on Nov. 21, 2007, and both of which are fully incorporated herein by reference.

BACKGROUND

In battery/power management applications, PMOSFETs (p-channel metal oxide semiconductor field effect transistors) can be used as high side switches for controlling charging and discharging loops. Since the PMOSFET is usually more expensive than an NMOSFET (n-channel MOSFET) with the same turn-on resistance, sometimes NMOSFETs are used as high side switches. If the NMOSFET is used as the high side switch in an electronic system, e.g., a battery operating system, a charge pump can be used in order to fully turn on the NMOSFET even when the battery voltage is relatively low. However, the voltage of the battery may vary during operation. Consequently, a driving voltage generated by the charge pump may be too high that may break down the NMOSFET when the battery voltage is relatively high. To solve this problem, an extra voltage clamp can be used to limit the driving voltage to a predetermined value. However, extra power will be consumed by the voltage clamp and therefore the power efficiency of the electronic system is degraded.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated in and form a part of this specification, illustrate embodiment of the present invention and, together with the description, serve to explain the principles of the invention:

FIG. 1 depicts an exemplary block diagram of an electronic system, in accordance with one embodiment of the present invention.

FIG. 2 depicts an exemplary block diagram of an electronic system, in accordance with one embodiment of the present invention.

FIG. 3 depicts an exemplary circuit diagram of a charge pump, in accordance with one embodiment of the present invention.

FIG. 4A depicts an exemplary circuit diagram of an electronic system with a clock generator, in accordance with one embodiment of the present invention.

FIG. 4B depicts an exemplary circuit diagram of an electronic system with a clock generator, in accordance with one embodiment of the present invention.

FIG. 5 depicts an exemplary circuit diagram of an electronic system with a clock generator, in accordance with one embodiment of the present invention.

FIG. 6 depicts an exemplary circuit diagram of an electronic system with a sensor, in accordance with one embodiment of the present invention.

FIG. 7 depicts an exemplary circuit diagram of an electronic system with a discharge path, in accordance with one embodiment of the present invention.

FIG. 8 depicts an exemplary flowchart of operations performed by an electronic system, in accordance with one embodiment of the present invention.

FIG. 9 depicts a block diagram of an electronic system, in accordance with another embodiment of the present invention.

FIG. 10 depicts a schematic diagram of an electronic system in FIG. 9, in accordance with one embodiment of the present invention.

FIG. 11 depicts a schematic diagram of an electronic system, in accordance with yet another embodiment of the present invention.

FIG. 12 depicts a flowchart of a method for controlling a charge pump system, in accordance with one embodiment of the present invention.

SUMMARY

In one embodiment, a circuit includes a charge pump and a feedback circuit. The charge pump coupled to a switch provides a control signal to the switch. The feedback circuit coupled to the charge pump receives the control signal and adjusts an operating frequency of the charge pump based upon the control voltage. The control voltage is adjusted to a predetermined target voltage by adjusting the operating frequency through the feedback circuit.

DETAILED DESCRIPTION

Reference will now be made in detail to the embodiments of the present invention. While the invention will be described in conjunction with these embodiments, it will be understood that they are not intended to limit the invention to these embodiments. On the contrary, the invention is intended to cover alternatives, modifications and equivalents, which may be included within the spirit and scope of the invention as defined by the appended claims.

Furthermore, in the following detailed description of the present invention, numerous specific details are set forth in order to provide a thorough understanding of the present invention. However, it will be recognized by one of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well known methods, procedures, components, and circuits have not been described in detail as not to unnecessarily obscure aspects of the present invention.

In one embodiment, a circuit includes a charge pump and a feedback circuit. The charge pump coupled to a switch provides a control signal to the switch. The feedback circuit coupled to the charge pump receives the control signal and adjusts an operating frequency of the charge pump based upon the control voltage. The control voltage is adjusted to a predetermined target voltage by adjusting the operating frequency through the feedback circuit.

FIG. 1 depicts an exemplary block diagram of an electronic system 100, in accordance with one embodiment of the present invention. As shown in FIG. 1, the electronic system 100 includes a charge pump driver 102 for generating an output 130 to control an electronic element 104. In addition, a clock generator 112 coupled to the charge pump driver 102 can generate a clock signal 114 to control the charge pump driver 102 and adjust a frequency f₁₁₄ of the clock signal 114 according to a status of the electronic element 104.

More specifically, in one embodiment, the electronic system 100 further includes a sensor 106 coupled to the electronic element 104 and for generating a control signal 110 indicative of the status of the electronic element 104. For example, the electronic element 104 can be a switch, e.g., an NMOSFET. The status of the switch 104, e.g., whether the switch 104 is fully turned on, can be indicated by a gate-source voltage V_(GS) of the switch 104. The sensor 106 can monitor the gate-source voltage V_(GS) of the switch 104, and generate the control signal 110 according to the gate-source voltage V_(GS). In addition, the switch 104 can be turned on by the output 130 of the charge pump driver 102. A voltage level of the output 130 can be pumped up by the charge pump driver 102 based on the clock signal 114.

In one embodiment, the control signal 110 can be an analog sensing signal V_(S) having a voltage level proportional to the gate-source voltage V_(GS). In another embodiment, the control signal 110 can be a digital control signal indicative whether the switch 104 is fully turned on. For example, the sensor 106 can generate the sensing signal V_(S) and compare the sensing signal V_(S) with a predetermined signal V_(PRE), and generate the digital control signal according to the comparison. In one such embodiment, the predetermined signal V_(PRE) is determined by a fully turn-on voltage V_(F) of the switch 104. When V_(S) is less than V_(PRE), the switch 104 is not fully turned on. When V_(S) is greater than V_(PRE), the switch 104 can be fully turned on.

Furthermore, the sensor 106 can control the clock generator 112 according to the control signal 110 indicative of the status of the switch 104, so as to adjust the frequency f₁₁₄ of the clock signal 114. For example, the frequency f₁₁₄ can have a first value f₁ when the switch 104 is not fully turned on. The frequency f₁₁₄ can also have a second value f₂ that is less than the first value f₁ when the switch 104 is fully turned on. The first value f₁ of the frequency f₁₁₄ can be, but is not limit to, higher than 500 KHz, such that the voltage level of the output 130 can be ramped up relatively quickly, so as to turn on the switch 104 relatively quickly. The second value f₂ can be, but is not limit to, ranged from 20 KHz to 500 KHz. As such, when the switch 104 is fully turned on, the charge pump driver 102 can be driven by the clock signal 114 with a lower frequency (e.g., from 20 KHz to 500 KHz), so as to maintain the voltage level of the output 130 to a substantially constant level and keep the switch 104 fully on. Thus, the power consumption can be reduced and the power efficiency can be increased. In one embodiment, the frequency f₁₁₄ of the clock signal 114 can be adjusted inversely proportional to the gate-source voltage V_(GS) of the switch 104.

FIG. 2 depicts an exemplary block diagram of an electronic system 200, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 1 have similar functions and will not be repetitively described herein.

The charge pump driver 102 can be used in battery charging and discharging systems. More specifically, a control switch 216 (e.g., a charging switch) and a control switch 218 (a discharging switch) can be controlled by the charge pump driver 102 according to the control signal 110 of the sensor 106. The charging switch 216 can be, but is not limit to, an NMOSFET. The discharging switch 218 can be, but is not limit o, an NMOSFET. The sensor 106 can be coupled to a gate 216G and a source 216S of the charging switch 216 to monitor a status, e.g., a gate-source voltage V_(GS), of the charging switch 216.

In one embodiment, the charging switch 216 and the discharging switch 218 can control the charging and/or discharging of a battery pack 242 according to outputs 230 and 232 of the charge pump driver 102. The battery pack 242 can be coupled to the charging switch 216. A terminal 240 of the discharging switch 218 can be coupled to a load (not shown in FIG. 2). In one embodiment, when the discharging switch 218 is turned on by the output 232 and the charging switch 216 is turned off by the output 230, the battery pack 242 can discharge power to the load via the body diode 234 of the charging switch 216 and the drain-source channel of the discharging switch 218. In an alternate embodiment, the charging switch 216 can be on during the battery discharging operation.

In another embodiment, the terminal 240 is coupled to a power source (not shown in FIG. 2). The power source can include, but is not limited to, an adapter or a universal serial bus device. In one such embodiment, when the charging switch 216 is turned on by the output 230 and the discharging switch 218 is turned off by the output 232, the battery pack 242 can be charged by the power source via the body diode 238 of the discharging switch 218 and the drain-source channel of the charging switch 216. In an alternate embodiment, the discharging switch 218 can be on during the battery charging operation.

In the example of FIG. 2, the charge pump driver 102 includes a non-overlapping clock generator 206, a charge pump 212 and a charge pump 214. The charge pumps 212 and 214 can be used to generate the outputs 230 and 232 to control the charging switch 216 and the discharging switch 218 respectively. The non-overlapping clock generator 206 can receive the clock signal 114 and generate a pair of complementary clock signals 226_1 and 226_0 for driving the charge pumps 212 and 214. In one embodiment, the charge pump driver 102 further includes a level shifter 208 and a level shifter 210. The level shifter 208 can be coupled between the non-overlapping clock generator 206 and the charge pump 212, and for converting the complementary clock signals 226_1 and 226_0 to a pair of complementary clock signals 222_1 and 222_0 having a voltage level V₂₂₂ that is higher than a voltage level V₂₂₆ of the clock signals 226_1 and 226_0. As such, the charge pump 212 can be driven by the pair of signals 222_1 and 222_0 with a higher voltage level V₂₂₂, e.g., V₂₂₂>V₂₂₆. Similarly, the level shifter 210 can convert the clock signals 226_1 and 226_0 to a pair of complementary clock signals 224_1 and 224_0 having a voltage level V₂₂₄ that is higher than the voltage level V₂₂₆, so as to drive the charge pump 214.

Furthermore, signals EN_CHG and EN_DSG can be used for enabling the charge pumps 212 and 214 respectively. In the example of FIG. 2, when the signal EN_CHG is enabled, the level shifter 208 can be enabled by the signal EN_CHG directly, and the OR-logic gate 236 can output a signal 228 to enable the non-overlapping clock generator 206 and the clock generator 112. As a result, the charge pump 212 can be enabled. The charge pump 214 can be enabled by the signal EN_DSG in a similar manner.

In operation, when the signal EN_CHG is disabled, the charge pump 212 can be disabled. A pull-down resistor 220 can be coupled between the gate 216G and the source 216S. When the charge pump 212 is disabled, a current flowing through the pull-down resistor 220 can decrease to zero, therefore the gate-source voltage V_(GS) can decrease to zero. As such, the charging switch 216 can be turned off.

In one embodiment, at the beginning of the battery charging operation when the signal EN_CHG is enabled, the sensing signal V_(S) is less than the predetermined signal V_(PRE), such that the control signal 110 can control the clock generator 112 to generate a higher frequency clock signal 114 (e.g., higher than 500 KHz). Thus, the charge pump 212 can be driven by the clock signals 222_1 and 222_0 with the higher frequency, and a voltage level of the output 230 of the charge pump 212 can be ramped up relatively quickly. The charging switch 216 can be turned on relatively quickly. Meanwhile, the voltage level of the sensing signal V_(S) can increase as the voltage level of the output 230 increases. When the sensing signal level V_(S) increases to the predetermined signal level V_(PRE), e.g., the charging switch 216 is fully turned on, the control signal 110 can control the clock generator 112 to generate a lower frequency clock signal 114 (e.g., range between 20 KHz and 500 KHz). Thus, the charge pump 212 can be driven by the clock signals 222_1 and 222_0 with the lower frequency, so as to maintain the voltage level of the output 230 at a substantially constant level and keep the charging switch 216 fully on.

Similarly, during a battery discharging operation, a separate sensor (not shown in FIG. 2) can be used to monitor the status of the discharging switch 218. The clock signal frequency f₁₁₄ can be adjusted according to the status of the discharging switch 218 in a similar manner as described with respect to the battery charging operation.

In one embodiment, the charging switch 216 can be the same as the discharging switch 218 and can be turned on during the battery discharging operation. In one such embodiment, the outputs 230 and 232 of the charge pumps 212 and 214 can have substantially the same voltage level. In addition, a voltage level V₂₄₀ at the terminal 240 can be lower than a voltage level V_(216S) at the source 216S when one of the switches 216 and 218 is not fully turned on, and can be substantially equal to the voltage level V_(216S) when both of the switches 216 and 218 are fully turned on. In other words, the gate-source voltage of the discharging switch 218 can be no less than the gate-source voltage of the charging switch 216. As such, when the charging switch 216 is fully turned on, the discharging switch 218 can be fully turned on as well. Thus, during the battery discharging operation, the frequency f₁₁₄ of the clock signal 114 for driving the discharging switch 218 can be adjusted according to the status of the charging switch 216.

Advantageously, the outputs 230 and 232 of the charge pumps 212 and 214 can be controlled by the clock signal 114 with an adjustable frequency f₁₁₄. As such, NMOSFETs with relatively low gate-source breakdown voltages can be used to further reduce the cost, in one embodiment.

FIG. 3 depicts an exemplary circuit diagram of a charge pump 212, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 2 have similar functions and will not be repetitively described herein. FIG. 3 is described in combination with FIG. 2.

As shown in FIG. 3, the charge pump 212 includes a first switch 302, e.g., an NMOSFET, coupled between a terminal 316 of a capacitor 318 and an input terminal 322 for charging the capacitor 318. The charge pump 212 can further include a second switch 314, e.g., a PMOSFET, coupled between the terminal 316 of the capacitor 318 and an output terminal 330 for discharging the capacitor 318. In addition, the output terminal 330 can be coupled to ground via a capacitor 320 and can provide the output 230 of the charge pump driver 102. The first and second switches 302 and 314 can be respectively controlled by the clock signals 222_1 and 222_0 which are generated according to the clock signal 114.

The charge pump 212 can also include a third switch 312, e.g., an NMOSFET, coupled between a terminal 306 of a capacitor 308 and the input terminal 322 for charging the capacitor 308. The charge pump 212 can further include a fourth switch 304, e.g., a PMOSFET, coupled between the terminal 306 of the capacitor 308 and the output terminal 330 for discharging the capacitor 308.

More specifically, the drain of switch 314, and the gates of switches 304 and 312 can be coupled to the terminal 316. The drain of switch 304, and the gates of switches 314 and 302 can be coupled to the terminal 306. When a voltage level V₃₀₆ at the terminal 306 is higher than a voltage level V₃₁₆ at the terminal 316, and a voltage difference between the levels V₃₀₆ and V₃₁₆ is greater than both of the threshold voltages of switches 302 and 304, the switches 302 and 304 can be turned on. Meanwhile, the switches 312 and 314 can be turned off. As such, the capacitor 318 can be charged by a power source (not shown in FIG. 3) at the terminal 322 via the switch 302, and the capacitor 308 can discharge power to the capacitor 320 via the switch 304. Similarly, when the voltage level V₃₀₆ is lower than the voltage level V₃₁₆, and the voltage difference between the levels V₃₀₆ and V₃₁₆ is greater than both of the threshold voltages of switches 312 and 314, the switches 312 and 314 can be turned on, and meanwhile the switches 302 and 304 can be turned off. As such, the capacitor 308 can be charged by the power source at the terminal 322 via the switch 312, and the capacitor 318 can discharge power to the capacitor 320 via the switch 314.

In one embodiment, in operation, when both clock signals 222_1 and 222_0 are at a low voltage level, e.g., 0V, the voltage levels V₃₀₆ and V₃₁₆ can be ranged from 0V to a voltage level V₃₂₂ at the input terminal 322. In one embodiment, when the clock signal 222_1 is low, e.g., at 0V, and the clock signal 222_0 is high, e.g., at a voltage level V_(H) that is higher than both of the threshold voltages of switches 302 and 304, the voltage V₃₀₆ at the terminal 306 can be increased by V_(H). In other words, the level V₃₀₆ can be higher than the level V₃₁₆, and the voltage difference between the levels V₃₀₆ and V₃₁₆ can be greater than the threshold voltages of switches 302 and 304. As such, the switches 302 and 304 can be turned on and the switches 312 and 314 can be turned off, such that the capacitor 320 can be charged by the capacitor 308 and the capacitor 318 can be charged by the power source at the terminal 322. Similarly, when the clock signal 222_1 is high, e.g., at the level V_(H), and the clock signal 222_0 is low, e.g., at 0V, the capacitor 320 can be charged by the capacitor 318 and the capacitor 308 can be charged by the power source at the terminal 322.

Advantageously, the clock signals 222_1 and 222_0 can be a pair of complementary clock signals provided by the non-overlapping clock generator 206 and the level shifter 208 as described for FIG. 2. As such, the capacitors 308 and 318 can be charged by the power source at the terminal 322 in an alternate fashion, and the charge stored in the capacitors 308 and 318 can be transferred to the capacitor 320 in an alternate fashion. In one embodiment, a voltage level at the output terminal 330 can increase as a frequency f₂₂₂ of the clock signals 222_1 and 222_0 increase. When the frequency f₂₂₂ reaches a certain level, the voltage level at the output terminal 330 can be substantially equal to V₃₂₂ plus V_(H).

FIG. 3 shows an exemplary structure of the charge pump 212. The output of the charge pump in the example of FIG. 3 can have an output voltage that is equal to the input voltage V₃₂₂ plus V_(H). However, the charge pump 212 can have other configurations. For example, by cascading the charge pump shown in FIG. 3 with another identical charge pump, e.g., by coupling the output terminal 330 to the input of the other charge pump, the charge pump can output an output voltage that is equal to the input voltage V₃₂₂ plus 2*V_(H). Similarly, by cascading N charge pumps together, the charge pump can output an output voltage that is equal to the input voltage V₃₂₂ plus N*V_(H). The charge pump 214 shown in FIG. 2 can have the similar structure as the charge pump 212.

A gate capacitor C_(216G) of the gate 216G (shown in FIG. 2) can be used as a charge reservoir for the charge pump 212. Furthermore, a gate voltage V_(216G) at the gate 216G can be used as a bulk bias voltage of the switches 304 and 314, e.g., substrates of the switches 304 and 314 can be coupled to the gate voltage V_(216G).

In one embodiment, a load of the charge pump 212 includes the gate capacitor C_(216G) of the gate 216G and the pull-down resistor 220 shown in FIG. 2. A resistance of the pull-down resistor 220 can be relatively high, such that a current flowing through the resistor 220 and the charge pump 212 can be relatively small. As such, the power consumption generated by the load of the charge pump 212 can be neglected. The power consumption of the charge pump 212 can be caused by the switching loss of switches 302, 304, 312 and 314. In one embodiment, the switching loss of the charge pump 212 can be reduced as the frequency f₂₂₂ of the complementary clock signals 222_1 and 222_0 is decreased. Advantageously, the charge pump 212 can operate at a lower frequency after the switch (e.g., the charging switch 216, the discharging switch 218) is fully turned on, so as to reduce the power loss for the charge pump 212, in one embodiment.

FIG. 4A depicts an exemplary circuit diagram of an electronic system 400 with a clock generator 112, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 1 have similar functions and will not be repetitively described herein.

As shown in FIG. 4A, the clock generator 112 includes a current controlled oscillator for generating the clock signal 114. More specifically, in one embodiment, the current controlled oscillator 112 includes a current source for generating a current I_(CH) to charge a capacitor 408 and for adjusting the current I_(CH) according to a status of a control switch 104 (or the charging switch 216, the discharging switch 218 shown in FIG. 2). For example, a first current source 404 can be coupled to a terminal 416 of the capacitor 408 and a second current source 402 can be coupled to the terminal 416 via a switch 406 (e.g., an NMOSFET, a PMOSFET). As such, when the switch 406 is on, the current I_(CH) can be provided by both of the current sources 402 and 404, and can have a first level I_(CH1). When the switch 406 is off, the current I_(CH) can be provided by the current source 404 and can have a second level I_(CH2) that is less than the first level I_(CH1). In one embodiment, when the control switch 104 is not fully turned on, the control signal 110 of the sensor 106 can turn on the switch 406, such that the current I_(CH) can have the first level I_(CH1). When the control switch 104 is fully turned on, the control signal 110 of the sensor 106 can turn off the switch 406, such that the current I_(CH) can have the second level I_(CH2).

In one embodiment, the clock generator 112 further includes a comparator 412 coupled to the terminal 416 of the capacitor 408. The comparator 412 can compare a voltage V₄₁₆ at the terminal 416 with a reference voltage V_(REF) and generate the clock signal 114 based on the comparison. Moreover, the clock generator 112 can include a discharging switch 410 coupled to the capacitor 408 and for discharging the capacitor 408 according to the clock signal 114.

For example, the discharging switch 410 is an NMOSFET having a drain coupled to the terminal 416 and a source coupled to another terminal 418 of the capacitor 408, e.g., coupled to ground. In addition, the comparator 412 can have a positive input terminal coupled to the voltage V₄₁₆, a negative input terminal coupled to the reference voltage V_(REF), and an output terminal coupled to a gate of the NMOSFET 410 and for providing the clock signal 114. As such, when the voltage V₄₁₆ is less than the voltage V_(REF), the switch 410 can be turned off by a low voltage signal 114. Meanwhile, the capacitor 408 can be charged by the charging current I_(CH) so as to increase the voltage V₄₁₆. When the voltage V₄₁₆ is greater than the voltage V_(REF), the comparator 412 can output a high voltage signal 114 to turn on the switch 410. Meanwhile, the capacitor 408 can be discharged by the switch 410, and the voltage V₄₁₆ will be decreased. As a result, the comparator 412 can generate the clock signal 114 by charging and discharging the capacitor 408 in an alternate fashion.

In one embodiment, the frequency f₁₁₄ of the clock signal 114 is determined by the charging current I_(CH), e.g., is proportional to the charging current I_(CH). As such, when the control switch 104 is not fully turned on, the charging current I_(CH) can have a level of l_(CH1) and the clock signal 114 can have a frequency value of f₁. When the control switch 104 is fully turned on, the charging current I_(CH) can have a level of I_(CH2) and the clock signal 114 can have a frequency value of f₂ that is less than f₁.

FIG. 4B depicts another exemplary circuit diagram of an electronic system 400′ with a clock generator 112, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 1 and FIG. 4A have similar functions and will not be repetitively described herein.

As shown in FIG. 4B, the discharging switch 410 is a PMOSFET having a drain coupled to the terminal 416 of the capacitor 408 and a source coupled to the terminal 418 of the capacitor 408. In one such embodiment, the terminal 418 can be coupled to a supply voltage V_(CC), the positive terminal of the comparator 412 can be coupled to the reference voltage V_(REF), and the negative terminal of the comparator 412 can be coupled to the voltage V₄₁₆. Similar to the embodiment of FIG. 4A, the comparator 412 in FIG. 4B can generate the clock signal 114 based on the comparison between the voltages V₄₁₆ and V_(REF).

FIG. 5 depicts another exemplary circuit diagram of an electronic system 500 with a clock generator 112, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 1 have similar functions and will not be repetitively described herein.

In the example of FIG. 5, the clock generator 112 includes a voltage controlled oscillator 510 for generating the clock signal 114. The clock generator 112 can further include a multiplex switch 502. The multiplex switch 502 can have a first terminal 508 coupled to the voltage controlled oscillator 510, a second terminal 504 coupled to a first reference voltage V_(REF1) and a third terminal 506 coupled to a second reference voltage V_(REF2). In addition, the multiplex switch 502 can be controlled according to the status of the control switch 104, e.g., according to the gate-source voltage V_(GS) of the control switch 104.

More specifically, the frequency f₁₁₄ of the clock signal 114 can be proportional to an input voltage, e.g., at the terminal 508, of the voltage controlled oscillator 510. The first reference voltage V_(REF1) can be greater than the second reference voltage V_(REF2). When the control switch 104 is not fully turned on, the control signal 110 of the sensor 106 can select the first reference voltage V_(REF1) as the input voltage of the voltage controlled oscillator 510, e.g., couple the terminal 508 to the terminal 504. As such, the voltage controlled oscillator 510 can generate the clock signal 114 with a higher frequency having the first value f₁. When the control switch 104 is fully turned on, the control signal 110 can select the second reference voltage V_(REF2) as the input voltage, e.g., couple the terminal 508 to the terminal 506. As such, the clock signal 114 can have a lower frequency having the second value f₂ that is less than f₁.

FIG. 6 depicts an exemplary circuit diagram of an electronic system 600 with a sensor 106, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 1 and FIG. 2 have similar functions and will not be repetitively described herein.

In the example of FIG. 6, the sensor 106 includes a sensing switch 602 having a first terminal 614 coupled to the control switch 216 and a second terminal 608 coupled to ground via a bias circuit 606 for providing the sensing signal V_(S). The sensing signal V_(S) can have a voltage level proportional to the gate-source voltage V_(GS) of the control switch 216. Furthermore, the sensor 106 can include a comparator 618 coupled to the clock generator 112. The comparator 618 can compare the sensing signal V_(S) with a predetermined signal V_(PRE), and generate the control signal 110 according to the comparison. In one embodiment, the predetermined signal V_(PRE) is determined by the fully turn-on voltage V_(F) of the control switch 216.

The sensing switch 602 can be a PMOSFET. The PMOSFET 602 can have a source, e.g., terminal 614, coupled to the gate 216G via a resistor 604, and a gate coupled to the source 216S and a drain, e.g., terminal 608, coupled to the bias circuit 606. As such, when the sensing switch 602 is cut off, a source-gate voltage V_(602SG) of the sensing switch 602 can be equal to the gate-source voltage V_(GS) of the control switch 216.

In one embodiment, a threshold voltage V_(T) of the sensing switch 602 can be equal to the fully turn-on voltage V_(F) of the control switch 216. As such, when the control switch 216 is not fully turned on (e.g., when V_(GS)<V_(F), V_(602SG)<V_(T)), the sensing switch 602 can be turned off and the sensing signal V_(S) at the terminal 608 can have a first voltage level V₁ that is lower than the predetermined signal level V_(PRE). When the control switch 216 is fully turned on (e.g., when V_(GS)≧V_(F), V_(602SG)≧V_(T)), the sensing switch 602 can be turned on, such that the sensing signal V_(S) can be ramped up by the gate voltage V_(216G) at the gate 216G and can have a second voltage level V₂ that is higher than the predetermined signal level V_(PRE). As a result, the sensor 106 can generate the sensing signal V_(S) indicative of the status of the control switch 216, and generate the control signal 110 to control the clock generator 112. In one embodiment, the bias circuit 606 includes a current source, e.g., a current mirror, that has a relatively high dynamic output resistance. In an alternate embodiment, the bias circuit 606 includes a resistor (not shown in FIG. 6) having a relatively high resistance.

In another embodiment, the threshold voltage V_(T) of the sensing switch 602 is less than the fully turn-on voltage V_(F) of the control switch 216. In one such embodiment, the sensing switch 602 can be turned on when the charge pump 212 is enabled. A current I₆₀₄ flowing through the resistor 604 can be given by I₆₀₄=V_(GS)/(R₆₀₄+1/g₆₀₂), where R₆₀₄ is a resistance of the resistor 604 and g₆₀₂ is a transconductance of the sensing switch 602. Thus, the current I₆₀₄ can be proportional to the gate-source voltage V_(GS) of the control switch 216. As shown in FIG. 6, the current mirror 606 can provide two substantially identical currents, e.g., a reference current I_(REF) and a mirror current I₆₁₀, when both MOSFETs 610 and 612 operate in the active (saturation) region. In one such embodiment, the level of the reference current I_(REF) can be properly chosen according to the fully turn-on voltage V_(F) of the control switch 216, such that the sense voltage V_(S) can indicate the status of the control switch 216.

More specifically, when the control switch 216 is not fully turned on (e.g., when V_(GS)<V_(F)), the current I₆₀₄ can be less than the reference current I_(REF). As such, the MOSFET 610 can drag the sense voltage V_(S) down to a first voltage level V′₁, e.g., lower than the predetermined signal level V_(PRE), so as to operate in the linear (ohmic) region and pass the current I₆₀₄ that is less than the reference current I_(REF). When the control switch 216 is fully turned on (e.g., when V_(GS)≧V_(F)), the current I₆₀₄ can be equal to the reference current I_(REF). Meanwhile, the MOSFET 610 can operate in the active (saturation) region, and sense voltage V_(S) can be at a second voltage level V′₂ that is higher than the predetermined signal level V_(PRE).

FIG. 7 depicts an exemplary circuit diagram of an electronic system 700 with a discharge path, in accordance with one embodiment of the present invention. Elements that are labeled the same as in FIG. 2 have similar functions and will not be repetitively described herein. FIG. 7 is described in combination with FIG. 2.

In one embodiment, the electronic system 200 in FIG. 2 can further include a first discharge path for disabling the charging switch 216 and a second discharge path for disabling the discharging switch 218. For example, the first discharge path includes a first switch 702 coupled between a first terminal, e.g., gate 216G, of the charging switch 216 and a second terminal, e.g., source 216S, of the charging switch 216. The first switch 702 can also coupled to ground via a second switch 704 for disabling the charging switch 216. The second discharge path can include a third switch 706.

More specifically, the first switch 702 can be a PMOSFET having a source coupled to the gate 216G of the charging switch 216, a gate coupled to the source 216S of the charging switch 216, and a drain coupled to the second switch 704. In addition, the second switch 704 can be controlled by a control signal 710.

During battery charging/discharging, the charge pump 212 can be enabled, and a voltage on the pull-down resistor 220 can be greater than both the threshold voltages V_(216T) of the charging switch 216 and V_(702T) of the first switch 702. As such, both of the switches 216 and 702 can be turned on. Meanwhile, the control signal 710 can be disabled so as to turn off the second switch 704.

In one embodiment, when the battery charging/discharging operation is disabled/terminated, the charge pump 212 can be disabled and the control signal 710 can be enabled. As such, the second switch 704 can be turned on and the first discharging path can be conducted. At the moment, the gate voltage V_(216G) at the gate 216G can decrease relatively quickly because a discharging current can flow through the first discharging path from the gate 216G to ground. In one embodiment, when the gate voltage V_(216G) decreases to a first level and the gate-source voltage V_(GS) of the charging switch 216 is less than the threshold voltage V_(216T), the switch 216 can be turned off. Similarly, when the gate voltage V_(216G) decreases to a second level and a source-gate voltage V_(702SG) of the switch 702 is less than the threshold voltage V_(702T), the switch 702 can be turned off. The first level can be equal to or different from the second level.

Consequently, in one embodiment, once the battery charging/discharging operation is disabled/terminated, the gate voltage V_(216G) can decrease relatively quickly, such that the charging switch 216 can be turned off relatively quickly. In addition, the first discharging path can be cut off when the gate voltage V_(216G) decreases to the second level, so as to avoid a current leakage flowing from the battery pack 242 (shown in FIG. 2) to ground.

The third switch 706 can be, but is not limit to, an NMOSFET or a PMOSFET, and can be controlled by a control signal 712. Similarly, during the battery charging/discharging, the charge pump 214 can be enabled, and the control signal 712 can be disabled so as to turn off the third switch 706. In one embodiment, once the battery charging/discharging operation is disabled/terminated, the charge pump 214 can be disabled and the control signal 712 can be enabled. As such, the switch 706 can be turned on so as to conduct the second discharging path. Consequently, a gate voltage V_(218G) of the discharging switch 218 can drop down to zero relatively quickly, such that the discharging switch 218 can be turned off relatively quickly.

FIG. 8 depicts an exemplary flowchart 800 of operations performed by an electronic system, in accordance with one embodiment of the present invention. FIG. 8 is described in combination with FIG. 1, FIG. 2, FIG. 3, FIG. 4A, FIG. 4B, FIG. 5.

In block 802, the clock generator 112 can generate the clock signal 114 to control the charge pump driver 102. The clock signal 114 can be generated by a current controlled oscillator (e.g., in FIG. 4A, in FIG. 4B). The clock signal 114 can also be generated by a voltage controlled oscillator (e.g., in FIG. 5).

In block 804, the clock generator 112 can adjust the frequency f₁₁₄ of the clock signal 114 according to a status of the electronic element 104, e.g., a switch. As such, the charge pump driver 102 can control the electronic element 104 based on the clock signal 114, as described in block 806. More specifically, the sensor 106 can generate the control signal 110 indicative of the status of the switch 104, e.g., whether the switch 104 is fully turned on or not. The sensor 106 can further control the clock signal 114 according to the control signal 110. For example, the frequency f₁₁₄ can be adjusted to the first value f₁ (e.g., higher than 500 KHz) when the switch 104 is not fully turned on. In addition, the frequency f₁₁₄ can be adjusted to a second value f₂ that is less than the first value f₁ when the switch 104 is fully turned on.

Accordingly, the present invention provides an electronic system for controlling an electronic element, e.g., a switch, according to a status of the electronic element. The electronic system can operate at a higher frequency to turn on the switch relatively quickly when the switch is not fully turned on, and operate at a lower frequency when the switch is fully turned on. In one embodiment, the electronic system can be used in battery charging/discharging applications, battery management systems, etc. The electronic system can be implemented in many other applications, such as mobile phones, laptops, digital cameras, portable media players, personal digital assistant devices, etc.

FIG. 9 illustrates a block diagram of an electronic system 900 according to one embodiment of the present invention. Elements labeled the same as in FIG. 2 have similar functions. The electronic system 900 includes a charge pump 910, a feedback circuit 920, an electronic device 930, a switch 940, and a power source 950. The switch 940 coupled to the power source 950 controls the connection of the power source 950 to the electronic device 930 based upon a control voltage V_(CTL). The charge pump 910 coupled to the switch 940 controls the control voltage V_(CTL) applied to the switch 940. The control voltage V_(CTL) controls a status of the switch 940, e.g., fully on, fully off, or partially on. The status of the switch 940 further determines a current flowing through the switch 940. In one embodiment, a discharging current flows through the switch 940 when the electronic device 930 is a load which is powered by the power source 950. In another embodiment, a charging current flows through the switch 940 when the electronic device 930 is a charger which charges the power source 950. In both circumstances, there are some considerations in controlling the control voltage V_(CTL). For example, the control voltage V_(CTL) is in a predetermined range for a normal charging/discharging process of the electronic system 900. Second, when the battery pack voltage B_(VAT) has a relatively large variation during operation, the control voltage V_(CTL) is avoided from being excessively high, since an excessively high control voltage V_(CTL) may break down the switch 940. Third, the control voltage V_(CTL) is avoided from being unnecessarily high, since an unnecessarily high control voltage V_(CTL) increases a power consumption of the electronic system 900.

To achieve the proper control voltage V_(CTL), the feedback circuit 920 receives the control voltage V_(CTL) and adjusts an operating frequency F_(CCO) of the charge pump 910 based upon the control voltage V_(CTL). By adjusting the operating frequency F_(CCO), the feedback circuit 920 adjusts the control voltage V_(CTL) to a predetermined target voltage V_(TARGET). As such, the proper control voltage V_(CTL) is achieved through the feedback circuit 920, which leads to advantages including, but not limited to safe charging/discharging process and improved power efficiency.

In one embodiment, the feedback circuit 920 includes a converter, e.g., a voltage-to-current converter (V/I converter) 904, and an oscillator 906. The V/I converter 904 generates a sense current I_(SEN) based upon the control voltage V_(CTL). In the example of FIG. 9, the feedback circuit 920 further includes a sensor 902 which is coupled to the switch 940 for generating a sense voltage V_(SEN) indicative of the control voltage V_(CTL). The V/I 904 coupled to the sensor 902 converts the sense voltage V_(SEN) into the sense current I_(SEN) based upon the sense voltage V_(SEN). The oscillator 906 coupled between the V/I converter 904 and the charge pump 910 outputs the operating frequency F_(CCO) to the charge pump 910 based upon the sense current I_(SEN). The feedback circuit 920 adjusts the operating frequency F_(CCO) of the charge pump 910 until the control voltage V_(CTL) is adjusted to the predetermined target voltage V_(TARGET).

FIG. 10 illustrates a schematic diagram of the electronic system 900 in FIG. 9 according to one embodiment of the present invention. Elements labeled the same as in FIGS. 2 and 9 have similar functions. In the example of FIG. 10, the battery pack 242 in FIG. 2 is used as the power source 950 in FIG. 9, and the discharging switch 218 in FIG. 2 is used as the switch 940 in FIG. 9. More specifically, the discharging switch 218 is an N channel metal oxide semiconductor field effect transistor (NMOSFET), in one embodiment. A positive terminal (PACK+) of the battery pack 242 is coupled to a drain of the discharging switch 218, and a negative terminal (PACK−) of the battery pack 242 is grounded. Furthermore, the charge pump 910 coupled to the battery pack 242 derives an output voltage from a voltage V_(BAT) of the battery pack 242. The output voltage is applied to a gate of the discharging switch 218. The load 930 is coupled between a source of the discharging switch 218 and ground.

The control voltage V_(CTL), e.g., a gate-to-source voltage V_(GS) in the example of FIG. 10, controls a status of the discharging switch 218. The charge pump 910 adjusts the control voltage V_(CTL) by adjusting the output voltage applied to the gate of the discharging switch 218.

In one embodiment, the sensor 902 includes a resistor 1002 and a resistor 1004, which are coupled in series between the gate and source of the discharging switch 218. In the example of FIG. 10, the voltage across the resistor 1002 is used as the sense voltage V_(SEN) indicative of the control voltage V_(CTL). The sense voltage V_(SEN) is given by

$\begin{matrix} {{V_{SEN} = \frac{V_{CTL}R_{a}}{R_{a} + R_{b}}},} & (1) \end{matrix}$ where R_(a) is the resistance of the resistor 1002, and R_(b) is the resistance of the resistor 1004.

In one embodiment, the V/I converter 904 includes a switch 1006 and a resistor 1008. For example, the switch 1006 is a PMOSFET. A gate of the switch 1006 is coupled to a conjunction node of the resistors 1002 and 1004, a source of the switch 1006 is coupled to the gate of the discharging switch 1002 through the resistor 1008, and a drain of the switch 1006 is coupled to the oscillator 906. When the sense voltage V_(SEN) is less than a reference voltage V_(REF), e.g., a threshold voltage V_(TP) of the switch 1006, the switch 1006 is turned off and no sense current is generated. When the sense voltage V_(SEN) becomes greater than the reference voltage V_(REF) (e.g., the threshold voltage V_(TP)), the switch 1006 is turned on and the sense current I_(SEN) is generated. A clamp voltage V_(CLAMP) can be predetermined according to:

$\begin{matrix} {V_{CLAMP} = {\frac{V_{TP}\left( {R_{a} + R_{b}} \right)}{R_{a}}.}} & (2) \end{matrix}$ Accordingly, when the control voltage V_(CTL) is greater than the clamp voltage V_(CLAMP), the V/I converter 904 generates the sense current I_(SEN). When the control voltage V_(CTL) is less than the clamp voltage V_(CLAMP), the sense current is cut-off. The sense current I_(SEN) can be given by

$\begin{matrix} {{I_{SEN} = {\frac{V_{SEN} - V_{TP}}{R_{S}} + {\frac{1}{2{KR}_{S}^{2}}\left( {1 + \sqrt{1 + {4{{KR}_{S}\left( {V_{SEN} - V_{TP}} \right)}}}} \right)}}},} & (3) \end{matrix}$ where K is a constant process parameter of the switch 1006, and R_(S) is the resistance of the resistor 1008. When R_(S) is big enough, equation (3) can be simplified to:

$\begin{matrix} {{I_{SEN} \approx \frac{V_{SEN} - V_{TP}}{R_{S}}} = {\frac{V_{DIF}}{R_{S}}.}} & (4) \end{matrix}$ According to equation (4), the sense current I_(SEN) is approximately proportional to the differential voltage V_(DIF) between the sense voltage V_(SEN) and the reference voltage V_(REF) (the threshold voltage V_(TP)).

In one embodiment, the oscillator 906 includes transistors 1010 and 1012, a capacitor 1016, a current source 1014, a switch 1018, and a comparator 1020. In the example of FIG. 10, the transistors 1010 and 1012 are PMOSFETs. By coupling gates together and sources together, the transistors 1010 and 1012 constitute a CMOS current mirror. The current source 1014 coupled to a drain of the transistor 1010 provides a bias current I_(B). The capacitor 1016 is coupled to a drain of the transistor 1012. When the sense current I_(SEN) is generated by the V/I converter 904, the current flowing through the transistor 1010 is equal to I_(B) minus I_(SEN). Through the CMOS current mirror, a current equal to I_(B) minus I_(SEN) flows through the transistor 1012, which is used for charging the capacitor 1016. As a result, a charging voltage is produced across the capacitor 1016. The comparator 1020 compares the charging voltage with a reference voltage V_(REF1) to output the operating frequency F_(CCO) to the charge pump 910. The operating frequency F_(CCO) can be given by

$\begin{matrix} {{F_{CCO} = \frac{I_{B} - I_{SEN}}{C_{a}V_{H}}},} & (5) \end{matrix}$ where C_(a) is the capacitance of the capacitor 1016, and V_(H) is the comparator hysteresis of the comparator 1020.

The charge pump 910 performs a charge pumping action by switching between connecting and disconnecting the battery pack voltage V_(BAT) (a charging voltage) to/from a flying capacitor in the charge pump 910 (not shown in FIG. 10) at the operating frequency F_(CCO). As a result, the charge pump 910 generates the output voltage applied to the gate of the discharging switch 218. On the one hand, the control voltage V_(CTL) (e.g., gate to source voltage V_(GS)) of the discharging switch 218 increases as charge is accumulated gradually at the gate of the discharging switch 218 due to the charge pumping action. On the other hand, the control voltage V_(CTL) decreases as the accumulated charge is dissipated by the resistors 1002 and 1004. A balance point is achieved when the charge pump 910 accumulates the charge at a speed approximately equal to the charge relaxation through the resistors 1002 and 1004. The operating frequency F_(CCO) of the charge pump 910 at the balance point is referred to as a balancing frequency.

When the operating frequency F_(CCO) of the charge pump 910 is higher than the balancing frequency, the control voltage V_(CTL) increases because the charge accumulation is faster than the charge relaxation at the gate of the discharging switch 218. When the operating frequency F_(CCO) of the charge pump 910 is lower than the balancing frequency, the control voltage V_(CTL) decreases because the charge relaxation is faster than the charge accumulation at the gate of the discharging switch 218. When the charge pump 910 operates at the balancing frequency, the balancing is achieved and the control voltage V_(CTL) is maintained substantially equal to the predetermined target voltage V_(TARGET). The predetermined target voltage V_(TARGET) can be given by

$\begin{matrix} {{V_{TARGET} = {\frac{R_{S}\left( {R_{a} + R_{b}} \right)}{R_{S} + R_{a}}\left( {\frac{V_{TP}}{R_{S}} + {I_{B}\frac{1}{1 + \frac{C_{a}V_{H}}{V_{BAT}C_{CP}}}}} \right)}},} & (6) \end{matrix}$ where C_(CP) is the capacitance of the flying capacitor in the charge pump 910. Assuming that C_(a)*V_(H) can be neglected compared to V_(BAT)*C_(CP), the equation (6) can be simplified to:

$\begin{matrix} {V_{TARGET} = {\frac{R_{S}\left( {R_{a} + R_{b}} \right)}{R_{S} + R_{a}}{\left( {\frac{V_{TP}}{R_{S}} + I_{B}} \right).}}} & (7) \end{matrix}$

More specifically, the control voltage V_(CTL) is lower than the predetermined clamp voltage V_(CLAMP) in an initial stage of operation, such that the switch 1006 is turned off and no sense current is generated. In this instance, the oscillator 906 outputs a predetermined maximum operating frequency to the charge pump 910, in one embodiment. At the predetermined maximum operating frequency, the control voltage V_(CTL) increases at a relatively fast speed in order to turn on the discharging switch 218. When the control voltage V_(CTL) reaches the predetermined clamp voltage V_(CLAMP), the switch 1006 is turned on and the sense current I_(SEN) is generated. Due to the sense current I_(SEN), the operating frequency F_(CCO) is decreased, and therefore the control voltage V_(CTL) increases at a slower speed. Though the increasing speed slows down, the control voltage V_(CTL) is still increasing. Accordingly, the sense current I_(SEN) increases, and the operating frequency F_(CCO) decreases to further slow down the increasing speed of the control voltage V_(CTL) until the balance point is achieved when the operating frequency F_(CCO) drops to the balancing frequency. At the balance point, the control voltage V_(CTL) is maintained to the predetermined target voltage V_(TARGET).

Similarly, when the operating frequency F_(CCO) becomes lower than the balancing frequency, the operating frequency F_(CCO) is increased through the feedback circuit 920 as the control voltage V_(CTL) decreases, until the balance point is reached. In other words, the feedback circuit 920 automatically adjusts the control voltage V_(CTL) and the operating frequency F_(CCO) until the predetermined target voltage V_(TARGET) is achieved.

Advantageously, the control voltage V_(CTL) is maintained to the predetermined target voltage V_(TARGET) regardless of a variation in the battery pack voltage V_(BAT). Due to the constant control voltage V_(CTL), the normal charging/discharging process of the electronic system 900 and the safety of the discharging switch 218 are maintained. In addition, when the charge pump 910 operates at the balancing frequency, the electronic system 900 has a lower power consumption compared to operating at other frequencies.

FIG. 11 illustrates a schematic diagram of an electronic system 1100 according to another embodiment of the present invention. Elements labeled the same as in FIGS. 2 and 10 have similar functions. Besides the discharging switch 218 and associated charge pump 910, sensor 902, V/I converter 904, and oscillator 906, the electronic system 1100 further includes the charging switch 216 and an associated charge pump 1110, a sensor 1120, and a V/I converter 1130.

The charging switch 216 controls a charging process of the battery pack 242 when the electronic device 930 is a charger. As discussed in relation to FIG. 9, the electronic device 930 can also be a load, and the discharging switch 218 controls the discharging process of the battery pack 242. The charge pumps 1110 and 910 can share the same operating frequency F_(CCO) output from the oscillator 906. In one embodiment, the sensor 1120 includes series-coupled resistors 1102 and 1104 and is coupled between gate and source of the charging switch 216. The V/I converter 1130 includes a switch 1106 and a resistor 1108. The wiring connection between the switch 1106, e.g., a PMOSFET, the sensor 1120 and the resistor 1108, is similar to that of the switch 1006, the sensor 902 and the resistor 1008 as discussed in relation to FIG. 10. In one embodiment, ratings of the charging switch 216 and associated elements for controlling the V_(GS) voltage (control voltage V_(CTL1)) of the charging switch 216 are substantially similar to the ratings of the discharging switch 218 and associated elements for controlling the V_(GS) voltage the discharging switch 218, respectively. The drains of the switches 1106 and 1006 for outputting the sense currents are OR wired together at node 1112. Adapted to the two sense currents, the current source 1014 is configured to provide a doubled bias current 2I_(B).

In a similar manner, the sensor 1120 generates a sense voltage indicative of the control voltage V_(CTL1) for the charging switch 216, and the V/I converter 1130 generates a sense current indicative of a differential voltage between the sense voltage provided by the sensor 1120 and a reference voltage, e.g., a threshold voltage of the PMOSFET switch 1106. Based upon the sense current from the V/I converters 904 and the sense current from the V/I converter 1130, the oscillator 906 adjusts the operating frequency F_(CCO) until the balance point is achieved.

By way of example, FIG. 10 shows the discharging switch 218 and corresponding elements for controlling the control voltage applied to the discharging switch 218; and FIG. 11 shows the discharging switch 218 and the charging switch 216 and corresponding elements for controlling the control voltages applied to the discharging switch 218 and the charging switch 216 respectively. However, the invention is not so limited. The electronic system 1100 can be expanded to include more switches and corresponding elements for adjusting the control voltages applied to these switches. In a similar manner, the electronic system 1100 can be adjusted to the balance point to achieve the proper control voltage for each of these switches.

FIG. 12 illustrates a flowchart 1200 of a method for controlling a charge pump system according to one embodiment of the present invention. FIG. 12 is described in combination with FIG. 10. Although specific steps are disclosed in FIG. 12, such steps are examples. That is, the present invention is well suited for performing various other steps or variations of the steps recited in FIG. 12.

In block 1202, a charge pump provides a control voltage. In one embodiment, the charge pump 910 generates the output voltage applied to the gate of the discharging switch 218. In one embodiment, the control voltage V_(CTL) is the V_(GS) voltage of the discharging switch 218.

In block 1204, the control voltage controls a status of a switch. In one embodiment, the control voltage V_(CTL) controls the status of the discharging switch 218, such as fully on, fully off, and partially on.

In block 1206, an operating frequency of the charge pump is adjusted base upon the control voltage. In one embodiment, the feedback circuit 920 adjusts the operating frequency F_(CCO) of the charge pump 910 based upon the control voltage V_(CTL). More specifically, the operating frequency F_(CCO) is adjusted based upon the differential voltage V_(DIF) between the reference voltage V_(REF) and the sense voltage V_(SEN) indicative of the control voltage V_(CTL.)

In block 1208, the control voltage is adjusted to a predetermined target voltage based upon the operating frequency. In one embodiment, the control voltage V_(CTL) is adjusted to the predetermined target voltage V_(TARGET) through the feedback circuit 920 when the operating frequency F_(CCO) reaches the balancing frequency.

While the foregoing description and drawings represent embodiments of the present invention, it will be understood that various additions, modifications and substitutions may be made therein without departing from the spirit and scope of the principles of the present invention as defined in the accompanying claims. One skilled in the art will appreciate that the invention may be used with many modifications of form, structure, arrangement, proportions, materials, elements, and components and otherwise, used in the practice of the invention, which are particularly adapted to specific environments and operative requirements without departing from the principles of the present invention. The presently disclosed embodiments are therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being indicated by the appended claims and their legal equivalents, and not limited to the foregoing description. 

1. A circuit comprising: a first charge pump coupled to a first switch and operable for providing a first control voltage to said first switch; and a feedback circuit coupled to said first charge pump and operable for receiving said first control voltage and for adjusting an operating frequency of said first charge pump based upon said first control voltage, wherein said first control voltage is adjusted to a predetermined target voltage by adjusting said operating frequency through said feedback circuit, wherein said feedback circuit comprises: a sensor coupled to said first switch and operable for generating a sense voltage indicative of said first control voltage; a first converter coupled to said sensor and operable for generating a first sense current based upon said sense voltage, wherein said first sense current is proportional to a differential voltage between said sense voltage and a reference voltage; and an oscillator coupled to said first converter and operable for adjusting said operating frequency based upon said first sense current.
 2. The circuit of claim 1, further comprising: a second switch coupled to said first switch; a second charge pump coupled to said second switch and operable for providing a second control voltage to said second switch; and a second converter coupled to said second switch and operable for generating a second sense current based upon said second control voltage, wherein said oscillator adjusts said operating frequency based upon said first sense current and said second sense current, and wherein said first charge pump and said second charge pump share said operating frequency.
 3. The circuit of claim 2, wherein said oscillator adjusts said operating frequency based upon a sum of said first sense current and said second sense current.
 4. The circuit of claim 1, wherein said first converter includes a P channel metal oxide semiconductor field effect transistor (PMOSFET), and wherein said reference voltage is a threshold voltage of said PMOSFET.
 5. The circuit of claim 1, wherein said first sense current is cut off when said first control voltage is less than a predetermined clamp voltage.
 6. The circuit of claim 1, wherein said first charge pump operates at a predetermined maximum frequency when said first control voltage is less than a predetermined clamp voltage.
 7. The circuit of claim 1, wherein said first switch comprises an NMOSFET, and wherein said first control voltage is a gate-to-source voltage of said NMOSFET.
 8. A method comprising: providing a control voltage by a charge pump; controlling a status of a switch based upon said control voltage; generating a sense voltage indicative of said control voltage, wherein said sense voltage is proportional to said control voltage; generating a sense current based upon said sense voltage; adjusting an operating frequency of said charge pump based upon said sense current; and adjusting said control voltage to a predetermined target voltage by adjusting said operating frequency.
 9. The method of claim 8, wherein said sense current is proportional to a differential voltage between said sense voltage and a reference voltage.
 10. The method of claim 8, further comprising: cutting off said sense current when said control voltage is less than a predetermined clamp voltage.
 11. The method of claim 8, further comprising: controlling said charge pump to operate at a predetermined maximum frequency when said control voltage is less than a predetermined clamp voltage.
 12. The method of claim 8, wherein said switch comprises an NMOSFET, and wherein said control voltage is a gate-to-source voltage of said NMOSFET.
 13. A system comprising: a power source; a switch coupled to said power source and operable for controlling a connection of said power source to an electronic device based upon a control voltage; a charge pump coupled to said switch and operable for providing said control voltage; and a feedback circuit coupled between said switch and said charge pump and operable for adjusting an operating frequency of said charge pump based upon said control voltage to a balance frequency, wherein said feedback circuit comprises: a sensor coupled to said switch and operable for generating a sense voltage indicative of said control voltage; a converter coupled to said sensor and operable for generating a sense current based upon said sense voltage; and an oscillator coupled to said converter and operable for adjusting said operating frequency based upon said sense current.
 14. The system of claim 13, wherein said converter includes a PMOSFET, and wherein said sense current is proportional to a differential voltage between said sense voltage and a threshold voltage of said PMOSFET.
 15. The system of claim 13, wherein said sense current is cut off when said control voltage is less than a predetermined clamp voltage.
 16. The system of claim 13, wherein said charge pump operates at a predetermined maximum frequency when said control voltage is less than a predetermined clamp voltage.
 17. The system of claim 13, wherein said switch comprises an NMOSFET, and wherein said control voltage is a gate-to-source voltage of said NMOSFET.
 18. The system of claim 13, wherein said sense voltage is proportional to said control voltage. 